Tuning capacitance to enhance FET stack voltage withstand

ABSTRACT

An RF switch to controllably withstand an applied RF voltage Vsw, or a method of fabricating such a switch, which includes a string of series-connected constituent FETs with a node of the string between each pair of adjacent FETs. The method includes controlling capacitances between different nodes of the string to effectively tune the string capacitively, which will reduce the variance in the RF switch voltage distributed across each constituent FET, thereby enhancing switch breakdown voltage. Capacitances are controlled, for example, by disposing capacitive features between nodes of the string, and/or by varying design parameters of different constituent FETs. For each node, a sum of products of each significant capacitor by a proportion of Vsw appearing across it may be controlled to approximately zero.

CROSS-REFERENCE TO RELATED APPLICATIONS-CLAIMS OF PRIORITY

This is a divisional application of and commonly owned U.S. applicationSer. No. 11/796,522, filed Apr. 26, 2007, now U.S. Pat. No. 7,960,772,and entitled “Tuning Capacitance to Enhance FET Stack VoltageWithstand”, and the contents of U.S. application Ser. No. 11/796,522 isincorporated by reference herein in its entirety.

BACKGROUND

1. Field

The present disclosure relates to electronic integrated circuits(“ICs”), and more specifically to circuits comprised of stackedtransistor devices for switching high frequency signals.

2. Related Art

Most radios, cell phones, TVs, and related equipment today require an“RF switch” to control connections between various transmitter andreceiver circuits (“RF” is used generically herein to mean anyreasonably high frequency ac signal). FIG. 1 is a simplified schematicdiagram of a typical, simple two throw switch that may by used toswitch, for example, a single antenna 102 between a transmit signalsource 104 and a receive circuit 106. Switches S₁ 108, S₂ 110, S₃ 112and S₄ 114 are represented by a mechanical single pole single throwswitch symbol. Typically, the switches are controlled such that when S₁is “closed” or conducting at low impedance, S₂ is “open” or highimpedance. Because no switch is perfect, the node of a transmit/receiveswitch (such as S₁ 108 or S₂ 110) farthest from the antenna is typicallyshunted to circuit common to reduce the effects of signal leakagethrough such switch when it is open. Thus, as S₂ 110 is illustrated inthe “open” state, the corresponding shunt switch S₄ 114 is “closed” toterminate the Receive S_(RF) signal on node 106 to ground 116.Conversely, the shunt switch S₃ 112 is “open” condition because itscorresponding signal switch S₁ 108 is “closed” to conduct the TransmitS_(RF) signal on node 104 to the antenna 102. To couple the antenna 102to the receive circuit, the condition of all four switches wouldtypically be inverted to that shown in FIG. 1.

In modern circuits, RF switches such as represented in FIG. 1 are mostoften implemented using semiconductor devices, typically some form offield effect transistor (FET). Semiconductor RF switches are commonlyfabricated using insulated gate FETs, often generically called MOSFETsdespite the fact that many do not employ the originalmetal/oxide/semiconductor construction that gave rise to that acronym.Non-insulating gate FETs, such as junction FETs (JFETs), are alsocommonly used, particularly with certain semiconductor materials such asGaAs. Each switch may be implemented using a single FET, or, asdescribed herein, a multiplicity of FETs stacked in series.

The impedance of ON (conducting) switches is generally sufficiently lowthat the voltage developed across it in this condition is negligible.However, switches that are OFF (nonconducting, or high impedance) musttypically support the full voltage of the RF signal they control. Thus,the RF power that can be controlled by a semiconductor RF switch dependson its voltage withstand capacity, which in turn depends on thedrain-to-source breakdown voltage (BVds) of its constituenttransistor(s). In FIG. 1, both S₂ 110 and S₃ 112 must withstand thetransmit signal voltage S_(RF) with respect to ground.

Integrated circuit fabrication requires many compromises. In particular,many IC transistors that are otherwise highly effective for switching RFsignals have a modest BVds, and thus may be inadequate to controlsignals of substantial amplitude. One solution may be to employalternative transistor designs yielding higher BVds. However, thetradeoffs necessary to fabricate transistors with higher BVds in anintegrated circuit may be burdensome. For example, such design may beincompatible with other circuitry desired for the integrated circuit, orit may otherwise be uneconomical.

Therefore, many semiconductor RF switches today stack a multiplicity oflow BVds transistors in series to improve the breakdown performance ofthe overall switch. FIG. 2 represents an example of such astacked-transistor semiconductor switch. The switch is disposed betweena first node N₁ 202 and a second node N₂ 204, and is controlled by avoltage V_(Control) 206. To form the overall switch, a multiplicity j ofFETs are “stacked” in series connection, from drain to adjacent source.Thus, a first transistor M₁ has a source coupled to N₁ 202, and a draincoupled directly to the source of a second FET M₂ 210. Additional FETs,represented by a series of dots, may be similarly connected above M₂210, the drain of the last such intervening FET being coupled to thesource of the top or j^(th) FET of the stack, FET M_(j) 212. Each FET ofthe stack is controlled by V_(Control) as coupled to the FET's gate viaa corresponding gate impedance, such as the base resistances RB₁ 214,RB₁ 216, . . . , RB_(j) 218 that are illustrated.

Though the FET channel terminal closer to N₁ is referred to as the“source,” and the opposite terminal as the “drain,” this is not arequirement. FETs may be implemented in a wide variety of designs andpolarities (e.g., N channel FETs and P channel FETs; enhancement anddepletion modes, and various threshold voltages, etc.). Moreover, thecircuits in which transistors are employed may be illustrated usingdifferent conventions than are followed herein. Transistor polarity anddrain-source orientation may often be interchanged without significantlyaltering the principle of operation of a circuit. Rather than illustratethe numerous possible permutations of drawing conventions, transistorpolarities, and transistor designs, it should be well understood bythose skilled in the electronics arts that the exemplary description andfigures illustrated herein equally represent all such alternativecircuit descriptions and equivalent device designs.

For most RF switch purposes, base impedances (represented in FIG. 2 asresistors RB_(x) 214, 216, 218) should combine with the effectivecorresponding gate capacitance of the FET to form a low-pass filterwhose transfer function has at least a single pole roll-off at afrequency that is less than ⅙ the lowest (expected) design frequency forthe signal that will exist between N₁ and N₂. Indeed, the at least onepole frequency is preferably 1/10 such lowest design signal frequency,or even lower. Such low frequency base control permits the gate voltageof each FET to follow the voltage on the channel of the FET, thusassuring the correct “on” or “off” gate/source voltage (Vgs), and alsolimiting both Vgs and the drain/gate voltage (Vdg) to prevent breakdownof the gate insulation.

Ideally, stacked device switches such as shown in FIG. 2 have a netvoltage withstand capacity equal to BVds of the individual FETs,multiplied by the number (j) of FETs in the stack. Thus, a stack of 10transistors each having BVds of 1.8 V would ideally be capable ofswitching a signal having a peak amplitude of 18 V. In practice,unfortunately, such stacks may be unable to support such ideal voltage.The voltage withstand capacity can be increased by increasing the numberof devices in the stack, but this may cause large increases in thecorresponding required integrated circuit area.

For example, assume that BVds for a given fabrication process is 2V(i.e., each single transistor can handle 2V), but that a 16V signal mustbe controlled. A stack of eight transistors should ideally be able tocontrol a signal of peak amplitude 16V. If eight transistors proveinsufficient for this task in practice, then more transistors must beadded to support the required voltage. Unfortunately, the seriesresistance of the stack is the sum of the individual device resistances.Consequently, as the number of stacked devices increases by a factor S,so does the ON resistance of the switch. Therefore, to maintain therequired overall ON resistance (or insertion loss), the impedance ofeach device must be reduced by the factor. S. This, in turn, requiresthat the area of each such device is increased by the factor S. Given Sadditional FETs, each with an area increased by S, it is clear that thetotal area of the FETs in the stack will increase as S². At some point,the switch area can be immense. Moreover, the parasitic capacitances ofthese transistors typically increase with the area, and this can lead tonumerous additional problems.

Thus, there is clearly a need to identify and solve the problem thatprevents some stacked FETs from controlling the ideal voltage, i.e., thenumber of FETs times the BVds of the individual FETs. Embodiments ofdevices and methods of manufacturing such devices are described hereinthat can mitigate or eliminate the noted problem, thus enabling stackedtransistors to withstand voltages that approach, or even equal, thetheoretical maximum for a given BVds of the constituent transistors.

SUMMARY

Research into observed failures of some stacked transistor RF switchesat lower than expected applied switch voltages (Vsw) led to a conclusionthat small parasitic capacitances (Cpd), previously thought negligible,were unexpectedly causing significant imbalances in the distribution ofVsw across the individual transistors of the stack. To reduce thedistribution imbalances, capacitance to internal stack nodes is added orintentionally modified, in contrast to the previous practice of merelymaking drain-source capacitances (Cds) uniform for the series-connected(stacked) transistors.

One embodiment is a stacked transistor RF switch comprising amultiplicity of constituent transistors (e.g., FETs) of the stack allcoupled in series connection drain to source to form a series string forwhich internal nodes are those between adjacent transistors. Thisembodiment includes effective drain-source capacitance Cds that issignificantly different for one transistor than for another transistorof the stack. Their relative Cds values may differ by an amount of atleast 2%, 5%, or 10%, or by at least 0.5% between each of a majority ofpairs of constituent transistors, and/or so as to effectively tunecapacitances of the stack. Tuning is effective if a variance of themagnitude of Vds-off as distributed across all constituent transistorsincreases when the Cds values are made substantially more equal. Anembodiment may include a discrete capacitive element coupled to aninternal node of the series string, and/or may include transistorshaving different Cds values due to design differences, and may havedifferent Cds between a majority of pairs of transistors of the stack.

Another embodiment is also a stacked transistor RF switch comprising amultiplicity of constituent transistors (e.g., FETs) of the stack allcoupled in series connection drain to source to form a series string forwhich internal nodes are those between adjacent transistors. Thisembodiment includes a discrete physical capacitor element Ccomp thateffectively tunes capacitances of the transistor stack by being coupledto an internal node of the series string. Tuning is effective if avariance of the magnitude of Vds-off as distributed across allconstituent transistors increases when all Ccomp capacitor elements areremoved. Ccomp capacitors may be fabricated as metal-insulator-metal(MIM) capacitors, or may be any other distinct physical feature havingan impedance that is predominantly capacitive at the frequency (aprimary frequency) of a signal ordinarily switched by the RF switch.

A further embodiment is a method of fabricating an RF switch comprisinga multiplicity of series connected constituent transistors in a seriesstring for which internal nodes are those between each pair of adjacenttransistors, and includes a step of establishing significantly differentvalues for total effective drain-source capacitance Cds of differenttransistors in the stack. Significantly different values may be thosevarying by at least 2%, 5%, or 10%, or by at least 0.5% between each ofa majority of pairs of constituent transistors, and/or may be such as toeffectively tune capacitances of the stack. Tuning is effective if avariance of Vds-off as distributed across all constituent transistorsdue to a voltage Vsw applied across the RF switch would increase if theeffective drain-source capacitances were substantially more equal. Themethod may include an additional step of determining parasitic draincapacitances, other than drain-source capacitances, coupled to internalnodes in the series string, and may include a further step ofdetermining voltages of nodes to which the parasitic drain capacitancesare coupled as compared to voltages on endnodes of the RF switch. Themethod may include a further capacitance balancing step of establishingvalues of the capacitances coupled to a particular internal node of theseries string of stacked transistors and multiplying each suchcapacitance value by a number reflecting a proportion of Vsw appearingacross the capacitance in operation to voltage-weight the capacitance,such that a sum of such voltage-weighted capacitances is approximatelyzero for the particular node. The method may further include thusbalancing a node between each of a majority of adjacent transistorpairs, or even thus balancing a node between each adjacent transistorpair of the stack.

A yet further embodiment is a method of fabricating an RF switchcomprising a multiplicity of series connected constituent transistors ina series string for which internal nodes are those between each pair ofadjacent transistors, and includes a step of coupling a discretecapacitive feature, or alternatively at least two discrete capacitivefeatures, to one or more internal nodes of the stack. A discretecapacitive feature is a distinct element having an impedance that ispredominantly capacitive at the frequency of a signal that is ordinarilyswitched by the RF switch, and the added capacitive feature(s) may berequired to effectively tune capacitances of the stack, such that avariance of magnitude in Vds-off as distributed across all constituenttransistors due to an RF switch voltage Vsw applied across the RF switchincreases when all such discrete capacitive features are omitted. Themethod may include an additional step of determining parasitic draincapacitances Cpd that are coupled to an internal node in the seriesstring, and also may include determining voltages of nodes to which theparasitic drain capacitances are coupled as compared to voltages onendnodes of the RF switch. The step of determining parasitic draincapacitances may include analyzing semiconductor device layout geometricparameters including parameters descriptive of interconnection traces.The method may include a capacitance balancing step of weighting eachvalue of the capacitances coupled to a particular internal node of theseries string of stacked transistors according to a number reflecting aproportion of Vsw appearing across the capacitance in operation, suchthat a sum of such weighted capacitance values is approximately zero forthe particular node. The method may further include performing thebalancing step for each of a majority of the internal nodes of theseries string, or for all of the internal nodes of the series string.

BRIEF DESCRIPTION OF THE DRAWINGS

Embodiments of the present invention will be more readily understood byreference to the following figures, in which like reference numbers anddesignations indicate like elements.

FIG. 1 is a simplified schematic diagram of a simple transmit/receive RFswitch.

FIG. 2 illustrates a basic FET stack designed to function as an RFswitch device.

FIG. 3 illustrates voltage division across j stacked FETs in an RFswitch that is “off.”

FIG. 4 illustrates effective parasitic drain capacitances Cpd for FETsin a stack such as illustrated in FIG. 2.

FIG. 5 is an equivalent circuit illustrating effects of a parasiticdrain capacitance Cpd.

FIG. 6 is an equivalent circuit illustrating an addition of tuningcapacitance between stack nodes to compensate Cpd in a circuit such asrepresented by FIG. 5.

FIG. 7 is a graph illustrating the relative Vds of each of thetransistors in a stack of 16 transistors as a function of a ratio of Cpdto Cds values.

FIG. 8 is a graph illustrating the effective number of transistors intransistor stacks having 2 to 16 transistors, as a function of a ratioof Cpd to Cds values.

FIG. 9 schematically illustrates further capacitance details of circuitssuch as illustrated in FIG. 4.

DETAILED DESCRIPTION

The background set forth above describes a typical stacked RF switch, asillustrated in FIG. 2. FIG. 3 illustrates how a switch such as shown inFIG. 2 should divide an applied RF voltage, such as (V_(S) 302-V_(Ref)304) such that equal voltages are imposed on each of the j FETs in thestack. When the switch is off (and presuming V_(Ref) is zero), V_(S) 302is applied to the endnodes 302-304 of the stack. Though each FET is in ahigh impedance state, the switch conducts somewhat due to the effectivedrain-source capacitances Cds corresponding to each FET_(X). Because the“off” conduction is almost entirely due to these capacitances, the FETstructure itself is not shown, but only the j corresponding effectivedrain-source capacitances Cds₁ 306, Cds₂ 308, Cds₃ 310, . . . ,Cds_((j-1)) 312 and Cds_(j) 314. This capacitive divider distributes theimpressed voltage V_(S) across each corresponding drain node to produceVd₁ 316, Vd₂ 318, Vd₃ 320, . . . , Vd_((j-2)) 322 and Vd_((j-1)) 324.

If each Cds_(x) has the same value, it appears that V_(S) should divideuniformly across the FETs such that Vd₁ (all voltages with respect toV_(Ref)) is V_(S)/j, Vd₂ is 2*V_(S)/j, and so on, withVd_((j-1))=(j−1)V_(S)/j. Because of this expected result, stacked FETdevices have previously been fabricated to establish Cds for each FET atsubstantially the same value. Other parasitic capacitances in such FETstacks are generally quite small relative to Cds, and moreover typicallydo not directly divide the impressed voltage. Accordingly, the effectsof such other parasitic capacitances have been largely ignored in regardto voltage division across the FETs in stacked-FET RF switches.

The series coupling of transistors to form a stacked switch asrepresented, for example, in FIG. 2 forms a path for conduction when theswitch is “on.” In that “on” condition, the conduction path couplestogether the endnodes (N₁ 202 on the “bottom” and N₂ 204 on the “top”)via the channels of all of the constituent FETs M₁ 208 to M_(j) 212.Only nodes along such conduction path will be referred to herein as“series nodes” or “nodes of the series string” of a transistor stack orstacked switch. Typically, most of the series string nodes are either adrain or a source of a constituent FET, which are so closely coupledthat all such nodes may be called “drain nodes.” However, other elementsmay be disposed in the series string of a transistor stack, and if somay have nodes that are on the conduction path, and that are thereforealso “series nodes” or “nodes of the series string.”

Voltage Distribution Inequality Due to Parasitic Drain Capacitances

After examination of a problem of unexpectedly low voltage breakdown forstacked-FET RF switches, the Applicant determined that voltagedistribution across the FETs in such stack was not uniform. Thereforeone FET generally experienced a larger proportion of the total appliedswitch voltage than any other FET in the stack. That most heavilystressed FET failed first, leading to failure of the other FETs in adomino-like effect. Upon further investigation, the Applicant determinedthat the voltage distribution inequalities are often caused by parasiticcapacitances that are small compared to the drain-source capacitances,and hence have been previously overlooked.

As noted above, drain parasitic capacitance (Cpd) values are generallynot more than a few percent of the corresponding drain-sourcecapacitance Cds. As such, Cpd has often been ignored when calculatingthe expected voltage distribution. Even small values of Cpd, however,may have large effects on the voltage distribution, depending upon thenode to which such Cpd is coupled. A Cpd coupling a drain to a node Npthat experiences some combination of the signals at the RF switchendnodes may inject such signals into the drain, changing thedistribution of voltages across the FETs of the stack from the idealequality to something less desirable. That is, if the node Npexperiences a signal that comprises (A*V₁+B*V₂), where A and B arepossibly complex or even time-varying multipliers, V₁ is the voltage atone end of the “open” RF switch, and V₂ is the voltage at the other endof the open RF switch, then a signal will be injected into thecorresponding drain that distorts the distribution of (V₂−V₁) across theFETs. Any Cpd that couples to such an Np (a node having a significantcomponent of (A*V₁+B*V₂)) may be relevant to net voltage distribution.When the signal on such Np differs greatly from the “ideal” drainvoltage, the signal injection and resulting voltage distributioninequality can be quite substantial, even if Cpd is small, due to thelarge voltage across it. This effect is described in more detail belowwith reference to FIG. 4.

Not all integrated circuit technologies have been equally successful inenhancing RF switch breakdown voltage by stacking devices in series. Insome technologies, Cpd may be quite obviously significant in comparisonwith a corresponding Cds, and hence would not be overlooked. However,the problem has also not been recognized or resolved with thesetechnologies, possibly because the performance has been so bad that theissue has not been pursued. Indeed, the very large parasitic componentsmight possibly be an important reason that stacked switch designs havenot been pursued using some integrated circuit technologies.

Special case: parasitic drain capacitances Cpd are predominantlydisposed between a corresponding drain and ground. In a special case,each significant Cpd_(n), for 1<=n<=(j−1), is coupled to a commonvoltage V_(COM) (e.g., ground) to which one end of the corresponding RFswitch is connected. Stated differently, each Np_(n) experiencesessentially the same signal as does one end of the RF switch, typicallyV_(COM) or ground. This special case is addressed first for two reasons:first, it approximates many practical switches; and second, it isconceptually simple.

An RF switch is often disposed between an RF signal node and a ground orcircuit common node. This is the situation, for example, for RF switchS₃ 112 of the transmit/receive antenna switch circuit illustrated inFIG. 1. As shown in FIG. 1, the transmit RF signal Transmit S_(RF) 104is imposed on one side of S₃ 112, while the other side of the switch S₃is connected to ground 116. It is also not unusual for parasitic draincapacitances Cpd to be predominantly coupled to a ground plane. Forexample, Cpd may predominantly consist of parasitic capacitances to asubstrate, and such substrate may be held at ground potential (at leastfor RF purposes). At least when both of these conditions exist, thespecial case arises in which each relevant Cpd is coupled to the signalat one end of the RF switch. In such special case, the Cpd of each FETwill have an effect on the (off) RF switch voltage distribution that isroughly proportional to the value of such Cpd multiplied by the number nof the FET, where n=1 for the FET whose source is coupled to V_(COM) orground, such that n indicates how many FETs are connected in seriesbetween the drain of FET_(n) and V_(COM) or ground.

FIG. 4 illustrates three FETs 402, 404 and 406 within a stack of j FETsthat are disposed between a first node N₁ 202 and a second node N₂ 204.For purposes of understanding the special case, it is temporarilyassumed that N₁ 202 is ground, and that Np_(n) 410 and Np_((n−1)) 414are also coupled to ground for at least RF signals. It is also presumedthat the ideal voltage on D_(n) (the drain of the center FET M_(n) 402)is (n/j)V_(N2). Such ideal voltage distribution obtains in the RF switchreflected by FIG. 4 if, for example, all values Cds are equal, and allvalues of Cpd are truly negligible. It is also momentarily assumed thatall values of Cds are indeed equal, and further that all values ofCpd_(x), for x unequal to n, are truly negligible.

The effect of Cpd_(n) 408 on the signal present on the drain D_(n) 502of FET_(n) may then be analyzed with reference to FIG. 5, which is anequivalent circuit for FIG. 4 that reflects the conditions andassumptions noted above. The equal-valued Cds capacitances for the (j−n)FETs above FET M_(n) are equivalent to a capacitor 504 having a value ofCds/(j−n). Similarly, the Cds for the lowest n FETs are equivalent to acapacitor 506 having a value Cds/n. As noted above, Cpd values haveconventionally been ignored because they are generally not more thanabout 2% of the values of Cds. Consideration of FIG. 5 reveals, however,that the effect of Cpd_(n) is much greater than suggested by its size inproportion to Cds, at least when n approaches j for large stacks. Forexample, let j=16, n=15, and Cpd_(n)=2% of Cds. Though Cpd_(n) is only2% as large as Cds, it is 30% as large as the equivalent capacitor(capacitor 506) that is parallel to Cpd_(n); as such, Cpd_(n) is clearlynot negligible. In fact, the resulting voltage at D_(n) 502 becomes(0.9202)N₂, rather than the ideal value (were Cpd_(n) not present) of (15/16)N₂ or (0.9375)N₂. Just a single Cpd₁₅, having a value only 2% thatof Cds, thus causes the drain-source voltage across M₁₆ to be (1.276/16)N₂ rather than ( 1/16)N₂, which is a 27.6% increase in Vds₁₆.Moreover, the effect is greatly increased when each D_(n) has acorresponding Cpd_(n) to ground, as illustrated in FIG. 7.

FIG. 7 is a graph showing the relative distribution of the drain-sourcevoltage Vdsn for each of the FETs (n=1 to 16) in a stacked RF switch, asa function of the size of Cpd with respect to Cds. Relative Vds is theVds for each FET_(n), compared to the voltage of N₂/j. Because j is 16in this example, Relative Vds is the Vds of the particular FET comparedto 1/16 of the RF switch voltage. A curve is provided for each of the 16FETs in the stack, which are labeled on the right side of the graph asspace permits.

As expected from consideration of FIG. 5, FIG. 7 shows that thedisparity between the ideal expected voltage across each FET_(n) is mostpronounced at the ends of the stack, i.e., for n=1 and for n=16; and themagnitude of the Vds_(n) increases for higher values of n. Indeed, FET₁₆experiences a relative voltage 200% larger than the ideal value(Relative Vds=2) when each Cpd is just 1.6% as large as Cds. Theassumptions for this figure are that each Cds has the same value, eachCpd has the same proportional value of Cds, and each Cpd is coupled to anode that is the RF signal equivalent to the voltage at the sourceconnection of FET M₁, e.g., ground.

FIG. 8 also reflects the effect of uncompensated Cpd capacitances as afunction of the ratio of Cpd to Cds. The Effective Stack Height is theactual withstand voltage of the RF switch in units of BVds, which isassumed the same for each stack transistor. The Effective Stack Heightfor stacks of j FETs are shown for actual stack heights j=1 (no stack atall) to 16 transistors. When the drain parasitic capacitances Cpd arevery small (0.0001, or 0.01% as large) compared to the drain-sourcecapacitances Cds, the stack operates nearly ideally, with a withstandvoltage of j multiplied by the BVds of each FET. Thus, when the Cpd/Cdsratio=0.0001, a stack of 13 FETs (j=13) behaves essentially like anideal stack of 13 devices, and hence it begins at an effective stackheight of 13. Each other trace similarly begins at an Effective StackHeight value equal to the actual stack height of the switch, so thetraces need no labeling. As the ratio of Cpd/Cds increases, theeffective stack height decreases, because uncompensated Cpd values causeunequal distribution of voltage over the FETs of the stack, decreasingmost rapidly for the largest stack (j=16). As the ratio of Cpd to Cdsincreases, the transistors no longer equally share the source voltage,and the top transistor in the stack, M_(j), typically experiencessignificantly more voltage than any other transistor. When M_(j) breaksdown, the remaining transistors follow suit in a domino effect, so theeffectiveness is limited by the voltage across K. For a stack of 16, theideal breakdown voltage of the stack would be 16×BVds, but at a Cpd/Cdsratio of only 1.6% (0.016), it will fail at 8×BVds, thus having aneffective stack height of 8.

For the special case, at least, it is thus clear that if Cpd values arenot taken into account the stack is likely to fail at a much lowervoltage than expected. The FET that is positioned furthest from theground connection of an RF switch will most likely fail first, at atotal RF switch voltage that may be a fraction of the ideal or expectedpeak voltage. Some solutions to this problem are set forth below for thespecial case, followed by a generalization of the problem andcorresponding general solutions.

Solutions for the Special Case

FIG. 6 is an equivalent circuit similar to that of FIG. 5, andillustrates a set of solutions that differ from each other according toa value of k. To compensate or tune a node D_(n) 502, a node D_((n+k))602 is selected based on factors such as ease of layout andeffectiveness of the node 602. Nodes of higher k may be more effective,as described below. As an example, if j=16 and n=10, k may be set to anyvalue from 1 to 6 (i.e., j−n). Selection of the node 602 results in acapacitor 604 composed of the series combination of those Cds that areabove D_((n+k)) 602. Capacitor 604 therefore has a value of Cds/(j−n−k),for k<(j−n). When k=(j−n), of course, there is no capacitor 604 becauseD_((n+k)) 602 is tied directly to N₂ 204. As in FIG. 5, a capacitor 506represents the series combination of the Cds of all FETs up to FET M_(n)and hence has a value of Cds/n. The capacitor 506 is coupled, togetherwith Cpd_(n) 408, to ground 116. Tuning is achieved by adding acompensation capacitor Ccomp_(n) 608 to compensate for the disruptiveeffects of Cpd_(n) 408.

In one conceptually simple solution, k=(j−n) so that D_((n+k)) 602 istied directly to N₂ 204. Perfect tuning of D_(n) 502 is then readilyachieved by making Ccomp_(n) 608 equal to Cpd_(n)*(n/k). This turns outto be the solution for any value of k. Such tuning is an iterativeprocess, because the node D_((n+k)) 602 will need to be subsequentlycompensated due to the effect of Ccomp_(n) 608. In this special case(Cpd components all effectively coupled to the source of M₁), solutionsof the form illustrated in FIG. 6 are most readily implemented by firstcompensating the node D₁ (Cdp₁), and then proceeding to compensate eachsuccessive drain node.

It may be useful in some embodiments to let k=1, such that Ccomp_(n) issimply disposed parallel to the channel of M_((n+1)). One advantage maybe the relative simplicity of disposing Ccomp_(n) between two nearbynodes. Another advantage may follow if the design of M₍₊₁₎ can bealtered such that the inherent capacitance Cds_((n+1)) is significantlyincreased. Changes to the layout and design of the transistors M₁ toM_(j) may reduce the size of the needed Ccomp capacitances, and may evenobviate a need for some discrete Ccomp capacitances.

On the other hand, if k>1 (i.e., if Ccomp is coupled to a drain of atransistor higher in the stack), then the actual capacitance requiredfor such Ccomp will generally be proportionally reduced as k increases.Note, however, that the breakdown voltage of such Ccomp mustcorrespondingly increase. To implement the special case solution whenk>1 requires discrete Ccomp capacitors to bridge a plurality of FETs. Inthe special case (i.e., when the various effective Cpd are predominantlycoupled to a node equivalent to the end of the RF switch that is coupledto the lowest transistor M₁), a k>=1 solution may be implemented bydisposing Ccomp capacitors between drain nodes D_(n) and a drainD_((m>n)). Drain D_((m>n)) may, for example, be a node equivalent to theend of the RF switch to which M_(j) is coupled.

The desirability of such coupling to drains of FETs that are more remotein the stack depends on the fabrication parameters and layout of thetarget RF switch. Factors tending to make such remote-node couplingdesirable include: a) a layout that lends itself to such connection,especially if the Ccomp layout create no further undesirable parasiticcapacitances; b) availability of a capacitor suitable for voltagesgreater than BVds; and c) a dearth of space available for suchcapacitors. Indeed, if the breakdown voltage BVc of a compensatingcapacitor is sufficiently high, it may be useful to couple thecompensating capacitor to that endnode of the RF switch that is oppositethe node to which the Cpd to be compensated is most closely coupled. Thecapacitance required for a compensation capacitor is proportional to1/m, where m is the number of series FETs across which such compensationcapacitor is coupled. This effect may permit compensation capacitorsdisposed across a plurality of FETs to occupy less die area, which isalmost always beneficial.

The best tuning embodiment thus depends, among other things, on theaccessibility of the various drain nodes, as well as on the suitabilityof capacitors that are compatible with the fabrication parameters, aswell as on the space available for such capacitors and whether or notthey can be fabricated atop other structures without adding die area. Iftuning creates difficulties in layout, it may be desirable to compensateless than perfectly, permitting j to increase slightly.

A further tuning solution applies to the special case (in which each Cpdis coupled to ground or N₁), and in a basic form uses compensationcapacitance only across single transistors (i.e., k=1). Conceptually,this further solution first compensates for Cpd₁ by increasing theeffective Cds₂ by an amount equal to Cpd₁. Next it compensates for Cpd₂by increasing the effective Cds₃ by an amount equal to Cpd₂, butincreased by a factor of 2 (i.e., 2 Cpd₂) because Cpd₂ is coupled acrosstwo transistors, M₁ and M₂. Furthermore, Cds₃ must be increased overCds₂, which has already been increased by the value of Cpd₁. All Cpdbeing equal, and all original Cds being equal, and for n>1, eacheffective Cds should be increased by an amount Ccomp_(n) determinedaccording to the following geometric progression:

$\begin{matrix}{{Ccomp}_{n} = {\sum\limits_{i = 1}^{n - 1}\;{i \times {Cpd}_{i}}}} & \left( {{Eqn}.\mspace{14mu} 0} \right)\end{matrix}$Though conceptually described as beginning with compensation of Cpd₁,note that the equation may be evaluated in any order. All compensationsmust be present at the time of fabrication, of course, and thus therecan really be no “order” of compensation.

Compensation or tuning can rarely be absolutely precise, and the valueof further precision certainly approaches zero for uncompensated Cpdvalues that are 0.01% of Cds or less. Smaller stacks tolerate greaterimprecision. In the case illustrated in FIG. 5, the single Cpd₁₅ thatwas presumed present resulted in a Vds₁₆ increased over the ideal valueby only a factor of 1.28. Yet FIG. 7 suggests that the result for thesame transistor, M₁₆, when each drain has a corresponding Cpd, would bethat Vds₁₆ have approximately a factor of 2.2 greater than the ideal.Thus, while a single parasitic capacitance may not be negligible, it isunlikely to cause severe voltage distribution imbalances by itself.Therefore, errors in tuning any one particular node may be unimportantif most of the nodes are reasonably well tuned.

Moreover, even imprecise tuning may substantially raise the voltagewithstand capacity of a stacked-transistor RF switch. For example, whenthe Cpd capacitances of a stacked-transistor RF switch design arepredominantly coupled to a first endnode of an RF switch, improvementsin switch voltage withstand may be realized by progressively increasingthe net effective Cds for transistors that are progressively fartherfrom the first endnode. Such general, progressive increase may beachieved, for example, by modifying transistor designs, and/or by addingdiscrete compensating capacitances. Such a general, imprecise solutionmay be as described in regard to FIG. 6, with k=1.

General Case Circuits and Solutions

In practice, parasitic capacitances from internal nodes can couple toany number of places. In a standard CMOS IC they may couple to thesubstrate. In SOI or GaAs devices they may couple to the package ormetal on the back of the part. In all types of devices, the parasiticcapacitances can also couple to nearby metal lines. Constituent Cpdcapacitances coupled to any node having a signal comprisingX*V_(N1)−Y*V_(N2) may limit the RF handling capability of large stacksto less than j×BVds.

Not only the effective Cpd of a drain node, but also the effective Cdsand/or the effective Ccomp may be comprised of a plurality of distinctconstituent capacitances. The constituents of an effective Cpdcapacitance may well be coupled to a multiplicity of different circuitnodes, as may constituents of Ccomp. Cds are coupled between particularnodes, but may still comprise a plurality of constituent capacitances.Consequently, the general case is far more complicated than the specialcase described above.

FIG. 9 expands upon a portion of FIG. 4 to illustrate such greatercomplexity. FIG. 9 illustrates M_(n) 402 of FIG. 4, together withCpd_((n−1)) 412, which is coupled to the source node S_(n) and to thecorresponding terminal node Np_((n−1)) 414, and the two endnodes N₁ 202and N₂ 204 of the RF switch. FIG. 9 illustrates an expansion of theeffective Cpd_(n) 408 of FIG. 4, or an expansion of an effectivecompensation capacitance Ccomp, or both. In the first case, Cn_(A) 902,Cn_(B) 904 and Cn_(C) 906, which are terminated at nodes 908, 910 and912 respectively, represent constituent capacitances of Cpd_(n) 408.Node 908 is the RF equivalent of the second end of the RF switch, N₂204, while Node 912 is the RF equivalent of the first end of the RFswitch, N₁ 202. Finally, node 910 is the RF equivalent of a differentdrain, D_(q). Cpd_(n) 408 represents the parallel combination of suchconstituent capacitances, so the total capacitance of Cpd_(n) 408 willbe the sum of the three values of Cn_(A), Cn_(B) and Cn_(C), presumingthere are no other significant Cpd constituents.

In this general case, the equivalent node Np_(n) 410 of FIG. 4 may wellnot be an actual node. However, in any event, it is a mathematicalequivalent node having an equivalent signal content based on the signalvoltages of N₂, N₁, and on the relative magnitudes of Cn_(A) 902, Cn_(B)904 and Cn_(C) 906.

It may be useful to determine whether the effective signal on theequivalent node Np_(n) 410 is closer to the signal on N₂ or the signalon N₁. The relevant signal components of the (net effective) Cpd_(n)will most often cause the signal to fall somewhere between the idealvoltage of D_(n) and the voltage of N₂, or else between the idealvoltage of D_(n) and the voltage of N₁. In the former case, D_(n) isproperly said to be more closely coupled to N₂ than to N₁, while in thelatter case D_(n) is properly said to be more closely coupled to N₁.With M₁ coupled to N₁ (as shown in FIG. 4), compensation of effectiveCpd_(n) that is more closely coupled to N₁ requires an increase incapacitance between D_(n) and one or more nodes that are more closelycoupled to N₂. The converse is also true: compensation of effectiveCpd_(n) that is more closely coupled to N₂ requires an increase incapacitance between D_(n) and one or more nodes that are more closelycoupled to N₁. For each drain node n, the effect of each constituent ofCpd may be calculated as described with respect to FIG. 6, and theeffects of all such constituents combined to determine the effectiveCpd_(n).

In an alternative view of FIG. 9, the capacitances Cn_(A) 902, Cn_(B)904 and Cn_(C) 906, instead of exclusively representing constituents ofa Cpd, represent constituents of both a Cpd and of a Ccomp. According toone example of this view, node 912 is N₁, and capacitor Cn_(C) 906comprises substantially all of Cpd_(n). Node 910 (D_(q)) is the nexthigher drain, D_((n+1)) (i.e., q=n+1), and consequently the Cn_(B) 904represents an increase in effective drain-source capacitanceCds_((n+1)). Cn_(B) 904 may, for example, be a discrete capacitor, or itmay reflect an increase in Cds resulting from design modification ofM_((n+1)). Further, it may reflect a combination of both means. Node 908is an RF equivalent to N₂, so Cn_(A) 902 may be a small discretecapacitor coupled between D_(n) and N₂. Capacitances 902 and 904 areconstituents of Ccomp_(n). Values of the capacitances 902, 904 and 906,together with any disparity between Cds_(n) and Cds_((n+1)) notencompassed by Cn_(B) 904, should be established to satisfy Eqn. 1, asdescribed below. Of course, Ccomp may have any number of constituentcapacitances; and, as described above for determining the effective Cpd,the effect of each Ccomp constituent may be individually determined andthen combined as the effective Ccomp_(n).

A general rule for tuning or compensation of a node m, which is thedrain of a stacked transistor, is set forth below. Each capacitance Cimis disposed between the node m and a different node i. Node m (inoperation, with the RF switch in high impedance or off state) will havea voltage Vm, and under the same conditions each other node i will havea voltage Vi. P is the total number of distinct capacitors tied to nodem. Based on calculation of charge injection to the node m, then, balance(and thus uniform voltage distribution) may be achieved by establishingthat:

$\begin{matrix}{0 = {\sum\limits_{i = 0}^{P - 1}\;{\left( {{Vi} - {Vm}} \right) \times {{Cim}.}}}} & \left( {{Eqn}.\mspace{14mu} 1} \right)\end{matrix}$

To the extent that Cds for the transistor immediately above node m andCds for the transistor immediately below node m are equal, they may beignored if the voltage is uniform on those two transistors, i.e., if[V(m+1)−V(m−1)]/2=Vm. However, if the Vds above and below node m are notgoing to be established to be equal in magnitude (and opposite in sign),then each Cds must be included in the calculation. Even if the Vds areequal, the Cds values should be included in the summation at least tothe extent that they are significantly unequal.

Precision is helpful up to a point, but as noted above, precision is notalways necessary to substantially improve voltage distributionimbalances across the transistors in a stacked RF switch. For someembodiments, it will suffice to observe that the Cpd are, on average,more closely coupled to N₁ than to N₂, and accordingly to establishvalues for Cds that are significantly increasing (e.g., increasing bymore than 0.03%) for a majority of the FETs, rather than beingsubstantially equal or varying randomly. Many embodiments of the devicesand methods described herein achieve compensation for undesirableparasitic capacitances in a stacked FET RF switch by adding compensatingcapacitances coupled between nodes of transistors in the stack,particularly between drain nodes (or source nodes, which areequivalent). Such compensating capacitances may cause adjacent FETs in astack to have significantly different net values of Cds, or mayestablish a compensating network of capacitance that is parallel to theseries Cds string of the FET stack.

Adding Compensation Capacitance:

Cds is described as an effective drain-source capacitance, and hereinmeans the total effective drain-source capacitance, net of intentionaland unintentional capacitances and effects, unless a different meaningis made clear. Net effective Cds may be changed, for example, by simplycoupling a predominantly capacitive feature between the drain and sourcenodes of a transistor in an RF switch stack. A predominantly capacitivefeature is a passive element having an impedance at the frequency of aswitched signal that is more capacitive than inductive or resistive. Ascircuit designers will understand, many structures may be fabricatedthat function as capacitors, and any such capacitor or predominantlycapacitive feature or element may constitute a compensation capacitor orcapacitance.

A compensation capacitance may include a difference between theintrinsic Cds of different constituent transistors of a stack, at leastto the extent that such difference is an intentional result of specificdesign variations between such transistors. A particular constituent FETof a stack may have a different layout, or otherwise be designed orfabricated differently, from another constituent FET to achieve adesired difference in Cds values between such FETs. The manner in whichCds is achieved is not important to the devices and methods describedherein; instead, any technique may be employed to establish satisfactoryeffective Cds values. Thus, any significant or intentional differencebetween the effective Cds of different transistors in a stack may fairlybe considered to represent compensation capacitance.

When modifying individual transistor designs to vary their effective Cdsis feasible, such modification may at least partly tune the capacitanceof a transistor stack. Such modification may be very elegant. However,the required differences in design may be tedious to implement, and alsomay be relatively difficult to reestablish when a circuit must bemodified for unrelated reasons. Nonetheless, such modification mayprovide some or even all of the compensation capacitance needed tosatisfactorily tune the capacitances of a stacked transistor RF switch.

The simplest design modification for varying effective Cds is a simplechange in device size. A larger device intrinsically has a larger valueof Cds, so physically larger transistors may be used when larger Cds isrequired. Indeed, intrinsic Cds may well be substantially proportionalto device size. In the special case in which Cpd capacitances arepredominantly coupled to one endnode (the “bottom”) of the RF switch,the transistors at the top of the stack require more compensationcapacitance. In that case, the higher n transistors may be madeprogressively larger, either in lieu of, or in addition to, addingdiscrete capacitance between nodes of the series string of the stack.

The general idea of varying transistor size to at least partiallysupplant a need for discrete compensation capacitors applies to thegeneral case of tuning stacked transistor switches. However, thefollowing analysis particularly applies to the special case of a stackas described above and represented by FIG. 4, with the assumption madeabove for FIGS. 5 and 6 that each Cpd is coupled to a node equivalent tothe lowest endnode (N₁, to which M₁ is coupled) of the RF switch. Thewidth W_(n) of each transistor M_(n), n>1, may be determined as followsfor the special case, with W₁ selected to establish overall switchresistance to satisfy performance requirements:

$\begin{matrix}{W_{n} = {W_{1}{\prod\limits_{i = 1}^{j - 1}\;\frac{{Cds}_{i} + {i \times {Cpd}_{i}}}{{Cds}_{i}}}}} & \left( {{Eqn}.\mspace{14mu} 2} \right)\end{matrix}$Because Cds is generally a linear function with transistor width, whileCpd will typically be non-linear, eqn. 2 cannot readily be furthersimplified, nor made precise. Ideally, a stack tuned in accordance withEqn. 2 will also satisfy the requirements of Eqn. 1, as set forth above.

Capacitances fabricated using gate insulation may be employed fortuning, though they may, for example, have a relatively low breakdownvoltage, or be nonlinear. Moreover, because parasitic capacitance isoften proportional to layout area, adding such compensating capacitorsto the side of a transistor may create further parasitic capacitance.This may make tuning an iterative process, because the solution to theproblem changes with each addition of compensation.

Metal-Metal (MIM) capacitance disposed on top of the switch transistoritself may be the best solution in some cases. Thus located, MIMcapacitors may not require extra die area, and typically add no extraparasitic capacitance to ground, at least. Moreover, establishing adesired capacitance with a MIM capacitor is relatively simple, andconsequently is likely to be easier to revise for subsequent designiterations, as compared to solutions based on modifying the transistordesign. MIM capacitors may also have higher breakdown voltages, and thusmay be amenable to being coupled between nodes m and i that areseparated by a plurality of transistors (i.e., k>1 with respect to FIG.6).

Identifying and Quantifying Effective Tuning

A voltage Vsw applied to an RF switch comprised of j constituent stackedtransistors is distributed across the constituent transistors of thestack. Deviation from uniformity in the distribution may be quantifiedas the variance V of the portion of Vsw appearing on each transistor,where Vds₁ for each transistor M_(i) resulting from Vsw is Vi, and

$V^{2} = {\left( {\sum\limits_{i = 1}^{j}\;\left( {{Vi} - {{Vsw}/j}} \right)^{2}} \right)/{j.}}$Useful tuning of a stacked transistor RF switch will result in a smallervariance V for the distributed voltages across the constituenttransistors of the stack. Random process variation will inevitably causesmall differences between the Cds of different transistors in a stack.However, because a design with no variation will be as perfect aspossible, such random variations should generally act to increase thevariance of the distributed transistor voltages. Accordingly,intentional tuning through control of Cds, on the one hand, and random,unintentional variations in the values of Cds on the other hand, may bedistinguished by a showing of whether reducing the variation in Cds in adevice or method (making the Cds of constituent transistors moreuniform) would decrease or increase the variance in voltage distributionacross the stack. The variance will increase as a result of reducing theCds variations that serve to effectively tune a stacked switch.

Variance in voltage distribution across constituent transistors cansimilarly distinguish distribution tuning capacitance that is coupled toan internal node of the series string of a stacked-transistor switch.Upon omission of the predominantly capacitive elements coupled tointernal nodes of the switch in an RF switch embodiment as describedherein, the variance of the voltage distribution will increase if theyare distribution tuning capacitances. Conversely, if capacitance hasbeen coupled to internal string nodes for purposes other than tuning toincrease voltage withstand capacity, then removing such capacitance willdecrease the variance of voltage distribution.

Random process variation in Cds values can be distinguished fromvariations implemented intentionally to increase voltage withstandtuning by the magnitude of the maximum Cds variations. Thus, for theconstituent transistors of a stack on a particular device, the largestCds will be very close to the smallest Cds if the deviation is merelydue to random process variations. For tuning a stack of j transistors, asize comparison (Cds(max)/Cds(min)−1) may be required to be at leastj/200, or at least j/100, or at least j/50. Irrespective of j, a tunedstack of transistors may be required to have a Cds(max) that exceedsCds(min) by at least 2%, at least 5%, or at least 10%, or at least 20%.Any of these limitations may be explicitly added to any claim of method,process or apparatus in order to distinguish incidental designs that arenot intended to be encompassed by such claim.

Differences in net effective Cds values between adjacent pairs oftransistors in a series stack may be required to be at least 0.5% foreach of a majority of such adjacent pairs. Alternatively, differences insuch net effective Cds values between adjacent pairs of transistors maybe compared to a total Cpd (not including Cds constituents) for theinternal node of the string that is between the transistors of suchpair. The Cds differences may then be required to exceed the total Cpdfor the node between them, for at least half, or for a majority, of suchadjacent pairs. The calculation may also be made by averaging, such thata sum of Cds differences between all adjacent transistor pairs isrequired to exceed a sum of the total Cpd for the node(s) between suchpairs.

Determining Cpd values

The integrated circuit designer is often faced with a need to evaluatecircuit parasitic elements, and any such technique may be employed toestablish the parasitic drain capacitances Cpd to nodes other than thecorresponding source, as well as the parasitic drain-source capacitanceCds. Complete circuit simulation based on the detailed parameters of theselected fabrication process and layout is ideal, if the simulationprogram is accurate and sufficient processing power is available tocomplete the task in a reasonable length of time. It is also possible tobuild a circuit, probe (measure) the distribution of the RF switchvoltage across the individual transistors of the stack, and to deducethe effective Cpd values from such measurements. However, as notedabove, a substantial improvement in RF switch voltage withstand capacitymay be achieved even without perfect compensation. Accordingly, lessdemanding techniques may be employed to estimate Cpd values.

One example of such a technique to estimate parasitic capacitance from anode to substrate is as follows:

$\begin{matrix}{{C = {L\; ɛ\left\{ {\frac{w}{h} + 0.77 + {1.06 \times \left\lbrack {\left( \frac{w}{h} \right)^{0.25} + \left( \frac{t}{h} \right)^{0.5}} \right\rbrack}} \right\}}},} & \left( {{Eqn}.\mspace{14mu} 3} \right)\end{matrix}$where w and

L are the width and length of the node, t is the thickness of the node,h is the height of the node above the ground plane, ε=ε₀×8.854 e−12 F/m,and ε₀ is the relative permittivity of the substrate material.

Numerous computer programs exist to aid the designer in estimatingparasitic circuit elements. Programs such as Medici, ADS/Momentum,FastCap, HFSS, and others are capable of 2D and 3D parasitic capacitanceestimation. These tools enable more accurate estimation of capacitanceto all other nodes in the vicinity of the node(s) being analyzed.

Conclusion

The foregoing description illustrates exemplary implementations, andnovel features, of related methods of tuning or compensating thecapacitances of stacked-transistor RF switches to alleviate lowbreakdown voltages for such switch that would otherwise result fromimbalances in the distribution of the overall RF switch voltage acrossthe transistors of the stack. It also describes implementations andnovel features of integrated circuit stacked-transistor RF switches,apparatus that employ capacitive tuning or compensation features toimprove net breakdown voltage compared to the absence of such features.The skilled person will understand that various omissions,substitutions, and changes in the form and details of the methods andapparatus illustrated may be made without departing from the scope ofthe invention. Because it is impractical to list all embodimentsexplicitly, it should be understood that each practical combination ofthe features set forth above (or conveyed by the figures) as suitablefor embodiments of the apparatus or methods constitutes a distinctalternative embodiment of an apparatus or method. Moreover, eachpractical combination of equivalents of such apparatus or methodalternatives also constitutes a distinct alternative embodiment of thesubject apparatus or methods. Therefore, the scope of the presentedinvention should be determined only by reference to the appended claims,as they may be amended during pendency of the application, and is not tobe limited by features illustrated in the foregoing description exceptinsofar as such limitation is recited, or intentionally implicated, inan appended claim.

The transistors in stacked-transistor RF switches as described hereinare preferably of an insulated-gate type, or are biased so as to conductno DC gate current. Even more preferably, the transistors are FETs,particularly those referred to as MOSFETs, though that includes manyFETs that are not fabricated with traditional Metal/Oxide/Semiconductorlayers, as was once implied by the name. The FETs have been described asif they are of N polarity (NMOS), but they could equally well be PMOS.Embodiments may employ non-preferred transistors, though they mayrequire circuit adjustments to deal with control-node DC current.

The circuits illustrated and described herein are only exemplary, andshould be interpreted as equally describing such alternatives as may bereadily seen to be analogous by a person of skill in the art, whether bypresent knowledge common to such skilled persons, or in the future inview of unforeseen but readily-applied alternatives then known to suchskilled persons.

All variations coming within the meaning and range of equivalency of thevarious claim elements are embraced within the scope of thecorresponding claim. Each claim set forth below is intended to encompassany system or method that differs only insubstantially from the literallanguage of such claim, but only if such system or method is not anembodiment of the prior art. To this end, each element described in eachclaim should be construed as broadly as possible, and should beunderstood to encompass any equivalent to such element insofar aspossible without also encompassing the prior art.

What is claimed is:
 1. A method of fabricating a stacked RF switch thatincludes a multiplicity of series connected constituent transistors in aseries string for which internal nodes are those between each pair ofadjacent transistors, the method comprising: coupling a discretecapacitive feature to one or more internal nodes of the stack, where adiscrete capacitive feature is a distinct element having an impedancethat is predominantly capacitive at a primary frequency of a signalordinarily switched by the RF switch, balancing charge injection for aparticular internal node by estimating values of all significantcapacitive elements coupled to the particular internal node, estimatinga voltage across each such significant capacitive element, andcontrolling capacitance coupled to the particular node as necessary toapproximately zero a sum of all such capacitance values, as multipliedto reflect the corresponding estimated voltage.
 2. The method of claim1, further comprising approximately balancing charge injection for eachof a majority of internal nodes of the RF switch stack.
 3. The method ofclaim 2, further comprising approximately balancing charge injection foreach internal node of the RF switch stack.